Constant operation speed logic circuit

ABSTRACT

A logic circuit includes a plurality of input terminals (IN 1 , IN 2 , . . . ), an output terminal (OUT), a load (L), and at least two driver circuits (D 1 , D 2 , . . . ). Each of the driver circuits includes a plurality of gates connected in series, each gate being driven by one of the potentials at the input terminals. In addition, the first gates (Q 11 , Q 22 , . . . or Q 11  &#39;, Q 22  &#39;, . . . ) of the driver circuit connected directly to the output terminal are driven by different potentials at the input terminals.

BACKGROUND OF THE INVENTION

1. Field of the Invention

The present invention relates to a logic circuit such as a NAND circuit or an NOR circuit. Such a logic circuit is often used in address decoders, a bit line control clock generator circuit, etc of a memory device.

2. Description of the Prior Art

A prior art logic circuit such as a NAND circuit or a NOR circuit comprises a load, a single driver circuit driven by the potentials at two or more input terminals, and an output terminal connected between the load and the driver circuit. The driver circuit has a plurality of gates which are driven by the potentials at the input terminals.

In the above-mentioned prior art, however, the presence or absence of charges at an intermediate node of the gates generates a fluctuation in operation speed. In this case, the operation speed is dependent upon the condition of input signals. As a result, the overall operation speed is substantially reduced.

SUMMARY OF THE INVENTION

It is an object of the present invention to provide a logic circuit in which the operation speed is constant under any condition of input signals.

According to the present invention, at least two driver circuits are provided. Preferably, the same number of driver circuits is provided as input terminals. The driver circuits are connected in parallel with each other. The gates of the driver circuits connected directly to the output terminal are driven by different potentials at the output terminals. Therefore, at least one of the driver circuits operates rapidly in response to changes of input signals so as to change the potential at the output terminal. As a result, no fluctuation in operation speed is generated under any condition of the input signals, and, in addition, the operation speed is increased.

The present invention will be clearly understood from the description as set forth below contrasting the prior art with the present invention and referring to the accompanying drawings.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a circuit diagram of a prior art logic circuit;

FIGS. 2A through 2D and FIGS. 3A through 3D are timing diagrams of the signals appearing in the circuit of FIG. 1;

FIG. 4 is a circuit diagram of another prior art logic circuit;

FIGS. 5A through 5D and FIGS. 6A through 6D are timing diagrams of the signals appearing in the circuit of FIG. 4;

FIGS. 7A through 7D, FIGS. 8A through 8D, FIGS. 9A and 9B, and FIGS. 10A and 10B are circuit diagrams of embodiments of the logic circuit according to the present invention;

FIG. 11 is a circuit diagram of a semiconductor device to which the present invention is applied;

FIG. 12 is a circuit diagram of a memory cell of FIG. 11;

FIG. 13 is a circuit diagram of the bit line precharging circuit of FIG. 11;

FIG. 14 is a circuit diagram of the bit line sensing circuit of FIG. 11;

FIG. 15 is a circuit diagram of the pre-decoders of FIG. 11;

FIG. 16 is a circuit diagram of the bit line control clock generator circuit of FIG. 11;

FIG. 17 is a circuit diagram of the address buffer of FIG. 11;

FIG. 18 is a circuit diagram of the circuit for generating the pre-clock signals PA_(i) (i=0 to 7) of FIG. 16;

FIGS. 19A through 19E are timing diagrams for explaining the operation of the circuit of FIG. 16;

FIGS. 20A and 20B are circuit diagrams of the other embodiments bit line control clock generator circuit of FIG. 11; and

FIGS. 21A through 21E are timing diagrams for explaining the operation of the circuits of FIGS. 20A and 20B.

DESCRIPTION OF THE PREFERRED EMBODIMENTS

In FIG. 1, which is a prior art two-input complementary metal-insulator-semiconductor (CMIS) NAND circuit, two P-channel transistors Q₁ ' and Q₂ ' are connected in parallel between a power supply V_(cc) and an output terminal OUT to form a load L. The transistors Q₁ ' and Q₂ ' are driven by the potentials at the input terminals IN₁ and IN₂, respectively. In addition, two N-channel transistors Q₁₁ and Q₁₂ are connected in series between a power supply V_(ss), having a lower potential than the power supply V_(cc), and the output terminal OUT to form a driver circuit D₁. The transistors Q₁₁ and Q₁₂ are also driven by the potentials at the input terminals IN₁ and IN₂, respectively. Note that the "apostrophe" on the references designates "P-channel type".

The operation of the circuit of FIG. 1 will now be explained with reference to FIGS. 2A through 2D and FIGS. 3A through 3D. As illustrated in FIGS. 2A and 2B, when the potential at the input terminal IN₁ remains at V_(cc) and, in addition, the potential at the input terminal IN₂ is changed from V_(ss) to V_(cc), the potential at the output terminal OUT is changed from V_(cc) to V_(ss), as illustrated in FIG. 2D. In this case, however, before the change of the potential at the input terminal IN₂, the transistors Q₂ ' and Q₁₁ are both turned on and the transistors Q₁ ' and Q₁₂ are turned off, so that a node N₁ connecting the transistor Q₁₁ to the transistor Q₁₂ is charged at V_(cc) -V_(th3), where V_(th3) is the threshold voltage of the transistor Q₁₁, as illustrated in FIG. 2C. As a result, even when the potential at the input terminal IN₂ is changed from V_(ss) to V_(cc) so as to turn off the transistor Q₂ ' and turn on the transistor Q₁₂, the potential at the output terminal OUT does not become V_(ss) until the node N₁ is discharged, resulting in a relatively large delay time t₁.

On the other hand, as illustrated in FIGS. 3A and 3B, when the potential at the input terminal IN₂ remains at V_(cc) and, in addition, the potential at the input terminal IN₁ is changed from V_(ss) to V_(cc), the potential at the output terminal OUT is also changed from V_(cc) to V_(ss), as illustrated in FIG. 3D. This operation is logically the same as the above-mentioned operation. In this case, before the change of the potential at the input terminal IN₁, the transistors Q₁ ' and Q₁₂ are turned on and the transistors Q₂ ' and Q₁₁ are turned off, so that the potential at the node N₁ is at V_(ss), as illustrated in FIG. 3C. Therefore, when the potential at the input terminal IN₁ is changed from V_(ss) to V_(cc), so as to turn off the transistor Q₁ ', and turn on the transistor Q₁₁, the potential at the node N₁ is changed with a relatively small delay time t₂, since it is unnecessary to discharge the node N₁.

Thus, a difference in speed between the operation as shown in FIGS. 2A through 2D and the operation as shown in FIGS. 3A through 3D, is generated. This difference is inherent in the construction of the series connection of the transistors Q₁₁ and Q₁₂.

The above-mentioned difference in operation speed similarly occurs in the case of a NOR circuit as illustrated in FIG. 4, which is a prior art two-input CMIS NOR circuit. In FIG. 4, two N-channel transistors Q₁ and Q₂ are connected in parallel between the power supply V_(ss) and the output terminal OUT to form a load L, while two P-channel transistors Q₁₁ ' and Q₁₂ ' are connected in series between the power supply V_(cc) and the output terminal OUT to form a driver circuit D₁. As illustrated in FIGS. 5A through 5D, when the potential at the input terminal IN₁ remains at V_(ss) and, in addition, the potential at the input terminal IN₂ changes from V_(cc) to V_(ss), it is necessary to charge the node N₁, having the potential V_(th2), where V_(th2) is the threshold voltage of the transistor Q₁₁ ', whereby a relatively large delay time t₁ is generated. However, as illustrated in FIGS. 6A through 6D, when the potential at the input terminal IN₂ remains at V_(ss) and, in addition, the potential at the input terminal IN₁ is changed from V_(cc) to V_(ss), it is unnecessary to charge the node N₁. As a result, a relatively small delay time t₂ is generated. Therefore, a difference in operation speed is generated between the operation as illustrated in FIGS. 5A through 5D and the operation as illustrated in FIGS. 6A through 6D.

In the present invention, the relatively large delay time t₁ is shortened so as to eliminate the above-mentioned difference in operation speed.

FIGS. 7A, 7B, and 7C are first, second, and third embodiments of the present invention, i.e., a two-input CMIS NAND circuit, a three-input CMIS NAND circuit, and a four-input CMIS NAND circuit, respectively. FIG. 7D is a fourth embodiment of the present invention, also a four-input CMIS NAND circuit.

In the embodiment of FIG. 7A, two driver circuits D₁ and D₂, connected in parallel, are provided. In the driver circuit D₁, a first N-channel transistor Q₁₁, connected directly to the output terminal OUT, is driven by the potential at the input terminal IN₁, while a second N-channel transistor Q₁₂ is driven by the potential at the input terminal IN₂. In the driver circuit D₂, a first N-channel transistor Q₂₁ connected directly to the output terminal OUT is driven by the input terminal IN₂ and a second N-channel transistor Q₂₂ is driven by the input terminal IN₁.

When the potentials at the input terminals IN₁ and IN₂ are V_(cc) and V_(ss), respectively, the transistors Q₂ ', Q₁₁ and Q₂₂ are turned on and the transistors Q₁ ', Q₁₂ and Q₂₁ are turned off. As a result, the potential at the node N₁ is V_(cc) -V_(th), and the potential at the node N₂ is V_(ss). Therefore, when the potential at the input IN₂ is changed from V_(ss) to V_(cc), the transistor Q₂ ' is turned off and the transistor Q₂₁ is turned on, so that the potential at the output terminal OUT is changed quickly from V_(cc) to V_(ss) under the influence of the discharged node N₂. However, when the potential at the input terminal IN₂ remains at V_(cc) and, in addition, the potential at the input terminal IN₁ is changed from V_(ss) to V_(cc), the potential at the output terminal OUT is also changed quickly from V_(cc) to V_(ss) under the influence of the discharged node N₁, in the same way as in FIG. 1. That is, in all cases, the operation time is t₂ as shown in FIG. 3D.

In the embodiment of FIG. 7B, three driver circuits D₁, D₂, and D₃, connected in parallel, are provided. The structure of the circuit of FIG. 7B is similar to that of the circuit of FIG. 7A. That is, first N-channel transistors Q₁₁, Q₂₁ and Q₃₁ of the driver circuits, connected directly to the output terminal OUT, are driven by the potentials at the input terminals IN₁, IN₂ and IN₃, respectively. Therefore, when two of the potentials at the input terminals IN₁, IN₂ and IN₃ remain at V_(cc) and the other potential remains at V_(ss), the potential at only one of the nodes N₁, N₂ and N₃ remains at V_(ss). For example, when the potentials at the input terminals IN₁ and IN₂ remain at V_(cc), and the potential at the input terminal IN₃ remains at V_(ss), the potential at only the node N₃ is V_(ss). As a result, when the potential at the input terminal IN₃ is changed from V_(ss) to V_(cc), the potential at the output terminal OUT is changed quickly from V_(cc) to V_(ss) under the influence of the discharged node N₃.

In the embodiment of FIG. 7C, four driver circuits, D₁, D₂, D₃ and D₄, connected in parallel, are provided. The structure of the circuit of FIG. 7C is also similar to that of the circuit of FIGS. 7A or 7B. That is, first N-channel transistors Q₁₁, Q₂₁, Q₃₁ and Q₄₁ of the driver circuits, connected directly to the output terminal OUT, are driven by the potentials at the input terminals IN₁, IN₂, IN₃ and IN₄, respectively. Therefore, when three of the potentials at the input terminals IN₁, IN₂, IN₃ and IN₄ remain at V_(cc) and the other potential remains at V_(ss), the potential at only one of the nodes N₁, N₂, N₃ and N₄ remains at V_(ss). For example, when the potentials at the input terminals IN₁, IN₂ and IN₃ remain at V_(cc), and the potential at the input terminal IN₄ remains at V_(ss), the potential at only the node N₄ is V_(ss). As a result, when the potential at the input terminal IN₄ is changed from V_(ss) to V_(cc), the potential at the output terminal OUT is changed quickly from V_(cc) to V_(ss) under the influence of the discharged node N₄.

In the embodiment of FIG. 7D, a four-input CMIS NAND circuit is illustrated, but only two driver circuits D₁ and D₂ are provided. In this case, the potentials at the input terminals IN₁ and IN₂ are more important than the potentials at the input terminals IN₃ and IN₄. That is, first N-channel transistors Q₁₁ and Q₂₁ of the driver circuits D₁ and D₂, connected directly to the output terminal OUT, are driven by the potentials at the input terminals IN₁ and IN₂, respectively. Therefore, when the potential at input terminal IN₁ or IN₂ remains at V_(ss) and the potentials at the other input terminals are V_(cc), the potential at the node N₁ or N₂ is V_(ss). Therefore, when the potential at the input terminal IN₁ or IN₂ is changed from V_(ss) to V_(cc), the potential at the output terminal OUT is changed quickly from V_(cc) to V_(ss). However, when the potential at the input terminal IN₃ or IN₄ is changed from V_(ss) to V_(cc), the potential at the output terminal OUT cannot be changed quickly from V_(cc) to V_(ss). Therefore, the circuit of FIG. 7D is useful only when to specific input conditions exist.

FIGS. 8A, 8B, 8C and 8D are fifth, sixth, seventh and eighth embodiments of the present invention, i.e., CMIS NOR circuits. Note that the embodiments of FIGS. 8A, 8B, 8C and 8D roughly correspond to the embodiments of FIGS. 7A, 7B, 7C and 7D, respectively. For example, the circuit of 8A is the same as the circuit of FIG. 7A, except the load L and the driver circuits D₁ and D₂ are reversed in respect with the output terminal OUT and the P-channel transistors of the load L are changed to an N-channel type and vice versa.

The operation of the circuits of FIGS. 8A, 8B, 8C and 8D are similar to the operation of the circuits of FIGS. 7A, 7B, 7C and 7D, respectively. For example, in the embodiment of FIG. 8B, when two of the potentials at the input terminals IN₁, IN₂ and IN₃ remain at V_(ss) and the other potential remains at V_(cc), the potential at only one of the nodes N₁, N₂ and N₃ remains at V_(cc). For example, when the potentials at the input terminals IN₁ and IN₂ remain at V_(ss), and the potential at the input terminal IN₃ remains at V_(cc), the potential at only the node N₃ is V_(cc). As a result, when the potential at the input terminal IN₃ is changed from V_(cc) to V_(ss), the potential at the output terminal OUT is changed quickly from V_(ss) to V_(cc) under the influence of the charged node N₃.

FIGS. 7A through 7D and FIGS. 8A through 8D are CMIS logic circuits. However, note that the present invention can be also applied to enhancement/depletion (E/D) type MIS logic circuits and conventional MIS logic circuit.

In an E/D MIS logic circuits such as an E/D MIS NAND circuit as illustrated in FIG. 9A or an E/D MIS NOR circuit as illustrated in FIG. 9B, the load L is constructed by a deletion-type transistor Q_(L1) (N-channel) or Q_(L2) ' (P-channel) in which the source and gate are connected to each other. On the other hand, in a conventional MIS logic circuit such as a NAND circuit as illustrated in FIG. 10A or an NOR circuit as illustrated in FIG. 10B, the load L is constructed by a enhancement-type transistor Q_(L3) (N-channel) or Q_(L4) ' (P-channel) in which the drain and gate are connected to each other.

The application of the circuits according to the present invention will now be explained with reference to FIG. 11.

In FIG. 11, a 64 Kbit static random access memory (RAM) is illustrated. References MCA1 and MCA2 designate memory cell arrays formed by a large number of memory cells MC as illustrated in FIG. 12; RAB row address buffers; CAB column address buffers; I/O input/output circuits; GEN a bit line control clock generator circuit; P-RD pre-decoders, as illustrated in FIG. 15; RD row decoders; WD word drivers; CD column decoders; BLL bit line precharging circuits as illustrated in FIG. 13; and BLC bit line sensing circuits as illustrated in FIG. 14.

In a highly integrated large-capacity memory device, each memory cell MC is very small in size. Accordingly, the pitch between adjacent rows is also extremely narrow. Therefore, it is difficult to place one decoder for each row. In order to overcome this, one row decoder is allocated to every two or four word lines. For example, as illustrated in FIG. 15, the pre-decoders P-RD determine the selection modes Φ₀₀, Φ₀₁, Φ₁₀ and Φ₁₁. In the pre-decoders P-RD, four NAND circuits NA₁, NA₂, NA₃ and NA₄ of FIG. 7A are used.

The logic circuits according to the present invention can also be applied to the bit line control clock generator circuit GEN of FIG. 11, which will now be explained in more detail. One bit line precharging circuit and one bit line sensing circuit are provided for each bit line pair. That is, during the read mode, the bit line precharging circuits BLL are controlled by a clock signal φ_(BP) from the bit line control clock generator circuit GEN to precharge the corresponding bit lines. After that, the bit lines become in a floating state. Next, one selected memory cell is connected to each bit line pair. As a result, the small difference in potential between the bit line pair is sensed and enlarged by the corresponding bit line sensing circuit BLC, which is controlled by a clock signal φ_(BC) from the bit line control clock generator circuit GEN.

FIG. 16 is one example of the bit line control clock generator circuit GEN of FIG. 11. In FIG. 16, the generator circuit GEN comprises a NOR circuit NO₁ of FIG. 8D, a NOR circuit NO₂ of FIG. 8C, a NAND circuit NA₅ of FIG. 7A, and a plurality of inverters. Row address pre-clock signals PA₀ through PA₇ correspond to external row address signals ADRS_(i) (i=0 to 7, FIG. 17), respectively. For example, when the row address signal ADRS₀ changes, the row address pre-clock signal PA₀ remains at a high level for a definite time. Otherwise, the signal PA₀ always remains at a low level. Such pre-clock signals PA₀ through PA₇ are obtained by the circuits of FIGS. 17 and 18. Note that CE of FIG. 17 designates a memory control signal. The circuits NO₁, NO₂ and NA₅ of FIG. 16 together comprise a circuit for detecting the change of row address signals.

Note that the NOR circuit NO₁ comprises only two loads, since, in this case, the signals PA₀ and PA₁ corresponding to the row address signals A₀ and A₁ are more important than the signals PA₂ and PA₃ corresponding to the row address signals A₂ and A₃.

As illustrated in FIG. 19A, when at least one of the row address signals PA₀ through PA₃ changes, the output of the NOR circuit NO₁ reaches a high level. Similarly, when at least one of the row address signals PA₄ through PA₇ changes, the output of the NOR circuit NO₁ reaches a high level. In all cases, the output of the NAND circuit NA₅ reaches a high level. As a result, the clock signal φ_(BP) changes as illustrated in FIG. 19C, and the clock signal φ_(BC) changes as illustrated in FIG. 19D. Therefore, the bit line sensing circuit BLC does not perform a sensing operation during the precharging operation mode and a predetermined time after the precharging operation is completed. As a result, the potentials at the bit lines change as illustrated in FIG. 19E.

Generally, the connections between the generator circuit GEN and the circuits BLL and BLC are formed by aluminum, which has a small delay time, while the word line WL is formed by polycrystalline silicon, which has a large delay time. As a result, the potential at the portion of a word line far from the word drivers WD changes slower than the portion of the word line near the word drivers WD. The change of the former is illustrated by a dot and dash line in FIG. 19B. In this case, the change of the potentials of the bit lines is illustrated by a dot and dash line in FIG. 19E. Therefore, it is necessary to delay the clock signals φ_(BP) and φ_(BC) in accordance with the delay of the change of the word line portion. For this purpose, the circuits of FIGS. 20A and 20B are added to the circuit of FIG. 16.

In the combination of the circuits of FIGS. 16, 20A and 20B, four kinds of bit line control clock signals φ_(BP1), φ_(BP2), φ_(BP3) and φ_(BP4) are generated for the four blocks B1, B2, B3 and B4 of the memory cell arrays MCA1 and MCA2, respectively. In addition, four kinds of bit line sensing clock signals φ_(BC1), φ_(BC2), and φ_(BC3) and φ_(BC4) are generated for the blocks B1, B2, B3 and B4, respectively.

For example, the circuit of FIG. 20A comprises four resistors R₁, R₂, R₃ and R₄ and four NAND circuits NA₁₁, NA₁₂, NA₁₃ and NA₁₄. However, these NAND circuits do not always require two driver circuits, since the sequence of the input conditions for the NAND circuits is definite. In FIG. 20A, when the potential of the signal φ_(BP) is changed from a low level to a high level and, accordingly, the potential of the signal φ_(BP) is changed from a high level to a low level, the potentials of the signals φ_(BP1), φ_(BP2), φ_(BP3) and φ_(BP4) rise simultaneously, as illustrated in FIG. 21C. However, when the potential of the signal φ_(BP) is changed from a high level to a low level and, accordingly, the potential of the signal φ_(BP) is changed from a low level to a high level, the signals φ_(BP1), φ_(BP2), φ_(BP3) and φ_(BP4) fall into the order illustrated by the dashed and dotted lines of FIG. 21C, due to the presence of the delay circuits.

Similarly, the potentials of the clock signals φ_(BC1), φ_(BC2), φ_(BC3) and φ_(BC4) change in the same way as those of the clock signals φ_(BP1), φ_(BP2), φ_(BP3) and φ_(BP4), except for the delay times of the fall of the potentials, as illustrated in FIG. 21D. Thus, the delay time of each block B1, B2 B3, and B4 of the memory cell arrays MCA1 and MCA2 can be compensated. In this case, note that it is not important that the potentials of the signals φ_(BP1) through φ_(BP4) rise simultaneously. In other words, the signals φ_(BC),i +1 also be obtained by shifting the signal φ_(BCi). Note that FIGS. 21A, 21B and 21E are the same as FIGS. 19A, 19B and 19E, respectively.

As explained hereinbefore, according to the present invention, no fluctuation in operation speed is generated regardless of the condition of the input signals. In addition, the operation speed is increased. 

We claim:
 1. A logic circuit operatively connected to receive input signals, comprising:a first power supply; a second power supply having a potential different from that of said first power supply; N input terminals, N being an integer greater than or equal to two, for receiving respective input signals; an output terminal; load means operatively connected between said first power supply and said output terminal; and N driver circuits, operatively connected in parallel between said second power supply and said output terminal and being cross-coupled to each other, for coupling said output terminal to said second power supply, each of said driver circuits comprising N gates operatively connected in series, each of said N gates being driven by one of the respective input signals of said N input terminals, when the voltage of one of the input signals on one of said N input terminals becomes different from that of the remaining input signals, the potential of only one of said N gates is changed, so that when the voltage of the input signal of said only one of said N gates is again changed, the voltage at the output terminal is quickly changed.
 2. A logic circuit as set forth in claim 1, wherein each of said N gates comprises an N-channel enhancement-type transistor, and wherein the potential of said first power supply is higher than the potential of said second power supply.
 3. A logic circuit as set forth in claim 2, wherein said load means comprises a plurality of P-channel enhancement-type transistors, respectively, operatively connected to said plurality of input terminals, each driven by one of the respective potentials of said plurality of input terminals.
 4. A logic circuit as set forth in claim 2, wherein said load means comprises an N-channel depletion-type transistor having a drain operatively connected to said first power supply, having a source and having a gate operatively connected to said source and said output terminal.
 5. A logic circuit as set forth in claim 2, wherein said load means comprises an N-channel enhancement transistor having a gate, having a drain operatively connected to said gate and said first power supply and having a source operatively connected to said output terminal.
 6. A logic circuit as set forth in claim 1, wherein each of said N gates comprises a P-channel enhancement-type transistor, and wherein the potential of said first power supply is lower than the potential of said second power supply.
 7. A logic circuit as set forth in claim 6, wherein said load means comprises a plurality of N-channel enhancement-type transistors, respectively, operatively connected to said plurality of input terminals, each driven by one of the respective potentials of said plurality of input terminals.
 8. A logic circuit as set forth in claim 6, wherein said load means comprises a P-channel depletion-type transistor having a drain operatively connected to said first power supply, having a source and having a gate operatively connected to said source and said output terminal.
 9. A logic circuit as set forth in claim 6, wherein said load means comprises a P-channel enhancement transistor having a gate, having a drain operatively connected to said gate and said first power supply and having a source operatively connected to said output terminal.
 10. A logic circuit, operatively connected to receive first and second input signals, having first and second power supplies, the potential of the first power supply being different from that of the second power supply, said logic circuit comprising:an output terminal; first and second input terminals operatively connected to receive the first and second input signals, respectively; first and second driver circuits, operatively connected to said first and second input terminals and to first and second power supplies, for coupling said output terminal to said second power supply, said first driver circuit comprising: a first P-channel transistor, operatively connected to said output terminal and said first input terminal, driven by the first input signal; a second P-channel transistor, operatively connected between said first transistor and said second power supply and operatively connected to said second input terminal, driven by the second input signal, said second driver circuit comprising: a third P-channel transistor, operatively connected to said output terminal and said second input terminal, driven by the second input signal; and a fourth P-channel transistor, operatively connected betwen said third transistor and said second power supply and operatively connected to said first input terminal, driven by the first input signal.
 11. A logic circuit, operatively connected to receive first and second input signals, having first and second power supplies, the potential of said first power supply being different from that of the second power supply, said logic circuit comprising:an output terminal; first and second input terminals operatively connected to receive the first and second input signals, respectively; first and second driver circuits, operatively connected to said first and second input terminals and said first and second power supplies, for coupling said output terminal to said second power supply, said first driver circuit comprising: a first N-channel transistor, operatively connected to said output terminal and to said first input terminal, driven by the first input signal; a second N-channel transistor, operatively connected between said first transistor and said second power supply and operatively connected to said second input terminal, driven by the second input signal, said second driver circuit comprising: a third N-channel transistor, operatively connected to said output terminal and said second input terminal, driven by the second input signal; and a fourth N-channel transistor, operatively connected between said third N-channel transistor and said second power supply and operatively connected to said first input terminal, driven by the first input signal. 